1. Field of the Invention
The present invention relates to insulated DC-DC converters that output DC power, and specifically relates to a DC power supply including a circuit for reducing semiconductor switching loss and a snubber circuit that suppresses a surge voltage generated in a rectifier circuit when a switching device is turned on.
2. Description of the Related Art
DC-DC converters are used when an unstable DC power source is stabilized, when a DC voltage is changed or when it is necessary to provide output of a DC power source that is electrically insulated from an input. Among them, in a DC power supply, an input and an output of which are electrically insulated from each other, a transformer used for the insulation can be downsized in proportion to an increase in used frequency.
Meanwhile, there is a limit on an increase in switching frequency because of heat generation due to switching loss in a semiconductor switch. Therefore, an identical example configuration in which a commutation circuit using a resonant circuit is provided to reduce switching loss is disclosed in Japanese Patent Laid-Open Publication 4-368464 (Patent Document 1) and “Comparative Analysis of Two Zero-Current Switching Isolated DC-DC Converters for Auxiliary Railway Supply,” (O. Deblecker, A. Moretti, and F. Vallee, SPEEDAM 2008) (Non-Patent Document 1).
FIG. 7 illustrates an example of a conventional resonant circuit described in these documents. Reference numeral 101 denotes an input DC power source, reference numeral 102 denotes a filter circuit including a filter reactor and a filter capacitor, reference numeral 103 denotes a resonant switch circuit, and reference numeral 104 denotes a gate controller that controls on/off of respective semiconductor switches.
An operation of the circuit in FIG. 7 will be described. Reference numerals Q1 to Q4 denote semiconductor switches included in an inverter circuit, to which free-wheel diodes D1 to D4 are attached, respectively. A primary winding of a transformer T is connected between a connection point a between the semiconductor switches Q1 and Q2 and a connection point b between the semiconductor switches Q3 and Q4, and a secondary winding of the same is connected to a connection point c and a connection point d of a bridge including rectifier diodes D5 to D8, via a resonance reactor Lz. An output of the bridge is provided to a load RL via the filter circuit 102.
The resonant switch circuit 103 is interposed between the output side of the bridge rectifier circuit and the filter circuit 102.
The gate controller 104 gives on/off instructions to the semiconductor switches Q1 to Q4 and a semiconductor switch Qz for resonant circuit control, respectively. While the semiconductor switches may be bipolar transistors, MOSFETs, thyristors, gate turnoff thyristors, IGBTs or the like, here, a description will be provided using IGBTs as a representative example.
FIG. 8 is a diagram illustrating temporal change in operational waveforms for describing the conventional example in FIG. 7. Here, Iab is a current flowing between the connection points a and b, Vab is a voltage between the connection points a and b, Iz is a current flowing in the resonant circuit, Vz is a voltage of opposite ends of a resonant capacitor Cz. Io is a current free-wheeling from the bridge rectifier circuit through the filter circuit 102 and the load RL.
A circuit operation will be described using the aforementioned diagram. When an on signal is provided from the gate controller 104 to each of the semiconductor switches Q1 and Q4, whereby the semiconductor switches Q1 and Q4 in the inverter circuit are in a conductive state. The current Iab flows to convey energy from the input DC power source 101 to the load RL.
When a signal to turn on the semiconductor switch Qz for resonant circuit control in the resonant switch circuit 103 is provided from the gate controller 104 to turn on the semiconductor switch Qz for resonant circuit control at a time t0 before turning off the semiconductor switches Q1 and Q4 in the inverter circuit, a charge current flows into the resonant capacitor Cz from the input DC power source 101. The current Iz is a serial resonant current between the resonance reactor Lz and the resonant capacitor Cz.
A current flowing in the semiconductor switches Q1 and Q4 is a sum of the current Id flowing toward the load and the resonant current Iz and increases in a sinusoidal form. At that time, a voltage is generated in the resonant capacitor Cz, and the voltage becomes a voltage higher than a secondary voltage of the transformer T.
At a time t1, the charging is completed, and the voltage reaches a maximum value. Subsequently, the resonant capacitor Cz start discharging, and a discharge current flows out to a path through a free-wheel diode D9 and the resonant capacitor Cz. Here, where a turn ratio of the transformer T is 1:1, the current Id in the filter reactor Ld flows so as to have a constant value that is a sum of the current Iab and the resonant current Iz, and thus, when the resonant current Iz increases, the current Iab decreases.
At a time t2, the resonant current Iz and the current Id are equal to each other, and thus, the current Iab is zero. The discharge progresses, and finally, at a time t4, the resonant capacitor Cz has completely discharged, and the resonant current Iz becomes zero. Meanwhile, the current Id flowing in the filter reactor Ld is continuous, and thus, at the point of time when the discharge current Iz from the resonant capacitor Cz becomes zero, the current Id is switched to the current Io flowing in the rectifier diodes D5 to D8. The continuity of the current Id is maintained as described above.
Upon a turnoff signal being transmitted from the gate controller 104 to each of the semiconductor switches Q1 and Q4 to turn the semiconductor switches Q1 and Q4 off at a time t3 between the time t2 and a time t4 when the resonant current Iz flows in the free-wheel diode D9 and the primary-side current Iab of the transformer is zero, the primary-side voltage Vab of the transformer becomes zero, and a voltage of a level that is equal to an input DC power source voltage E is applied to the semiconductor switches Q1 and Q4.
This is because a slight amount of excitation current remaining in the transformer free-wheels in a path of the diode D3, the input DC power source 101 and the diode D2. The current in the semiconductor switches Q1 and Q4 is substantially zero at the time t3, and thus, almost no switching loss occurs in the process of the turnoff.
Meanwhile, when the semiconductor switch Qz for resonant circuit control is turned on at the time to, the resonant current Iz is gradually increased by the resonance reactor Lz, and in a transient state of the turning-on, the resonant current Iz still has a small value and thus, there is only small switching loss.
Also, when the semiconductor switch Qz for resonant circuit control is turned off in a period in which the free-wheel diode D9 is conductive and the resonant current Iz is positive, the current in the semiconductor switch Qz for resonant circuit control is already zero, and thus, in the process of the turnoff, no switching loss occurs. However, loss caused by recovery occurs in the free-wheel diode D9 for the semiconductor switch Qz for resonant circuit control.
At a time t5, a turn-on instruction is provided from the gate controller 104 to each of the semiconductor switches Q2 and Q3, and turning-on of the semiconductor switches Q2 and Q3 is started. At this time, the current Id flowing in the filter reactor Ld is equal to the current Io free-wheeling in the rectifier diodes D5 to D8. At this time, the current Iab starts flowing through the resonance reactor Lz, and thus, cannot rapidly increase. Furthermore, the current Id can be regarded as constant and thus changes so that the current Id is a sum of the current Io and the current Iab. Accordingly, the current Iab increases by the amount of decrease of the current Io.
Therefore, in a transient state of the semiconductor switches Q2 and Q3 being turned on, almost no current flows. Therefore, there is only a small turning-on loss. The current Iab gradually increases, and at a time t6, becomes equal to the current Id, and the current Io becomes zero. In a subsequent half cycle from times t0′ to t6′, the counterpart arms (semiconductor switches Q2 and Q3) operates based on a principle similar to that described above.
In fact, from the time t5 to the time t6, when the free-wheeling current Io flows in the diodes, a voltage in an opposite direction is applied from the primary side of the transformer to the diodes, a phenomenon called recovery (reverse recovery) occurs in a set of the rectifier diodes D5 and D8 or a set of the rectifier diodes D6 and D7, and at the time t6, a surge voltage is generated. The surge voltage oscillates for a certain period of time due to a resonance phenomenon caused by an inductance of the recovery path and a junction capacitance of the rectifier diodes.
FIG. 9 illustrates voltage and current waveforms of the rectifier diode D6 at that time. While characteristics of recovery is determined depending on the nature of the diode, as illustrated in FIG. 9, in many cases, the voltage exceeds a withstanding voltage of the diode, resulting in breakage of the diode. Also, even though the voltage does not exceeds the withstanding voltage of the diode, high-frequency electromagnetic noise is generated depending on the temporal change (dv/dt) of the voltage at that time, and may adversely affect other apparatuses.
For a general, transformer-used, insulated DC-DC converter, such surge voltage is often a problem, and a countermeasure for the surge voltage is taken by providing a RC snubber for each of rectifier diodes. For other conventional examples, as illustrated in FIG. 10, Patent Document 1 discloses that a connection point g between a resonant capacitor Cz and a semiconductor switch Qz for resonant circuit control and a connection point h between a filter capacitor FC and a filter reactor Ld are connected via a snubber diode Ds to make a charging current flow in the resonant capacitor Cz when a surge voltage is generated, thereby suppressing such overvoltage. Other conventional examples of countermeasures for a surge voltage generated during recovery are disclosed in, e.g., Japanese Patent Laid-Open Publication No. 2006-352959 (Patent Document 2), Japanese Patent Laid-Open Publication No. 2009-273355 (Patent Document 3) and Japanese Patent Laid-Open Publication No. 2008-79403 (Patent Document 4).
The countermeasure disclosed in Patent Document 1 will be described below with reference to FIG. 10. FIG. 11 illustrates temporal change in operational waveforms in FIG. 10. The operation in FIG. 10 will be described with reference to FIG. 11. In FIG. 11, a waveform of a current Is flowing in a snubber diode Ds is added to the voltages and current waveforms illustrated in FIG. 8. Times t0 to t6 in FIG. 11 are similar to those in the case of FIG. 7 regardless of operation of the snubber diode Ds. At a time t5, a free-wheeling current Io flows in all rectifier diodes, and thus, a voltage between points e and f is almost zero.
Accordingly, charge in a resonant capacitor Cz is released through a free-wheel diode D9, and thus, a voltage at opposite ends of the resonant capacitor Cz is almost zero. At a time t5, semiconductor switches Q2 and Q3 are turned on, thereby a primary current Iab of a transformer gradually starts flowing, and thus, the free-wheeling current Io on the secondary side decreases.
At a time 6 when the current Io is zero, a voltage is generated between the points e and f on the secondary side of the transformer T, and the current Is starts flowing through the resonant capacitor Cz and the snubber diode Ds, Cz is charged until the time reaches a time t8, and a certain constant voltage, which is determined depending on the difference in potential between opposite ends of a filter reactor Ld in a filter circuit, is generated in the resonant capacitor Cz. The voltage of the resonant capacitor Cz is retained until the semiconductor switch Qz for resonant circuit control is turned on next time. The operation after the time t8 is similar to that in the case of FIG. 7 except a voltage Vz of the resonant capacitor Cz.
The snubber diode Ds forms a path for surge voltage suppression jointly with the resonant capacitor Cz. At the time t6 in FIG. 11, as in the circuit operation in FIG. 8, the rectifier diodes have a reverse recovery, resulting in a surge voltage being generated in the rectifier diodes, that is, between the points e and f. It is described that in this case, surge components can be absorbed by a path including the resonant capacitor Cz, the snubber diode Ds and the filter capacitor FC, thereby preventing generation of an excessive surge voltage.
The conventional example circuit illustrated in FIG. 10, in which the snubber diode Ds has been introduced, has three problems below. The first problem is that an amount of charge corresponding to the snubber current Is flowing in the resonant capacitor Cz remains in the resonant capacitor, resulting in reduction of a range of operation of the function that reduces turnoff loss.
The second problem is that the resonant capacitor Cz serves as both of two functions, i.e., a resonance capacitor and a snubber capacitor, and thus, two types of currents flow in the resonant capacitor Cz, increasing heat generated by the capacitor itself and thereby increasing the volume of the capacitor. The last problem is that where the resonant capacitor Cz becomes larger due to the aforementioned problem, the wiring inductance is increased, resulting in a decrease in the surge absorption function as a snubber circuit.
First, the first problem will be described. In an operation of the circuit illustrated in FIG. 10, if the voltage of the resonant capacitor Cz is not zero at the time t0 when the semiconductor switch Qz for resonant circuit control is turned on, that is, the resonant capacitor Cz has initially been charged as a result of absorbing a surge voltage will be described with reference to FIG. 12.
Solid lines in FIG. 12 indicate a case where the semiconductor switch Qz for resonant circuit control is turned on when the resonant capacitor Cz has not been charged, and dashed lines indicate a case where the semiconductor switch Qz for resonant circuit control is turned on when the resonant capacitor Cz has been charged. At a time to, the semiconductor switch Qz for resonant circuit control is turned on, thereby the resonant current provided by a resonance reactor Lz and the resonant capacitor Cz start flowing, and an amplitude of the current is small if the resonant capacitor Cz has initially been charged. In FIG. 12, I0m is an amplitude of the current Iz when the resonant capacitor Cz has not initially been charged, I1m is an amplitude of a current Iz when the resonant capacitor Cz has initially been charged. There is the relationship indicated by expression (1) below between I0m and I1m.I1m=(1−α)I0m  (1)
Here, α is a ratio of a voltage of the initially-charged resonant capacitor Cz to an output voltage of the rectifier diodes. I0m can be expressed by expression (2). However, Vef(t0) is an output voltage (between the points e and f) of the diode bridge at the time t0.
                              I                      0            ⁢            m                          =                                                            C                z                                            L                z                                              ⁢                                    V              ef                        ⁡                          (                              t                0                            )                                                          (        2        )            
If the resonant current Iz is insufficient, the current Iab cannot be lowered to around zero at a time t2. Subsequently, at a time t3, the semiconductor switches Q1 and Q4 on the primary side is turned off, resulting in large turnoff loss being generated in the semiconductor switches Q1 and Q4 compared to a case where the current is around zero.
FIG. 13 illustrates waveforms when the circuit that normally operates in FIG. 7 with the snubber diode Ds in FIG. 10 added thereto is put into actual operation. Merely the resonant capacitor Cz has a voltage (around ⅙ of a resonance peak voltage) in advance before operation of a resonance switch at a time t0, and there is no time t2 when the current Iab becomes around zero, and thus, switching loss occurs.
The occurrence of switching loss naturally lowers a drive frequency, providing a restriction on reduction in size of the transformer T. Also, introduction of the snubber diode Ds to the circuit capable of normally decreasing switching loss provides conditions for disabling the operation of the circuit, narrowing the range of an input voltage and/or an output current, that is, an operational range of the DC-DC converter.
In order to expand the narrowed operational range, it is necessary to increase an electrostatic capacitance of the resonant capacitor Cz so that I1m is larger than the value of the current Iab when the semiconductor switches Q1 and Q4 are turned off in FIG. 12. This is related to the second problem of heat generation of the resonant capacitor Cz. It should be noted that the resonance reactor Lz includes a leakage inductance of the transformer T and thus, it is difficult to reduce the resonance reactance Lz from the perspective of manufacturing.
Next, the second problem will be described. The resonant capacitor Cz is a circuit that also serves as a snubber capacitor. Thus, two types of currents, i.e., a resonant current and the snubber current Is, flow in the resonant capacitor Cz. In other words, the current Iz in FIG. 11 is a sum of the components of the current Iz, which the resonant current in FIG. 8, and the components of the current Is in the snubber circuit in FIG. 11, and thus, high-frequency components of the current Iz in FIG. 11 are larger than those of the current Iz in FIG. 8.
Heat generated by a capacitor increased in proportion to a square of an effective current value provided by an inner resistor and a square root of a frequency due to a skin effect. As in the present DC-DC converter, for a physical size (volume) of the resonant capacitor Cz that operates with a high frequency, heat generation is dominant rather than an electrostatic capacitance necessary for resonance. As described above, there is the problem of the heat generated by the resonant capacitor Cz being increased due to an increase in high-frequency components of the current Iz by the snubber current Is and the volume of the resonant capacitor Cz being also increased in proportion to the increase in the heat.
Furthermore, capacitors are elements sensitive to heat, and thus, are adversely affected in terms of lifetime if used at high temperature. Furthermore, an increase in electrostatic capacitance of the resonant capacitor Cz for expanding the operational range, which has been referred to as a solution to the first problem, leads to an increase in effective current value according to expression (2), which substantially increases the volume of the resonant capacitor Cz.
Lastly, the third problem will be described. When the heat generated by the resonant capacitor Cz is large, a space for cooling is required. In that case, the wiring inductance of the resonant capacitor Cz inevitably increases. In particular, such increase is significant in the case of large power. An effect of the wiring inductance on the snubber circuit that suppresses a surge voltage will be described.
At the time t6 in FIG. 11, a stepped voltage Vcd is generated at the output points e and f of the rectifier diode bridge, together with a surge voltage generated by recovery of the rectifier diodes. FIG. 14 illustrates an equivalent circuit of a path of a surge current flowing through the snubber circuit where such time is newly set to t=0 on a temporal axis and V2 is a voltage. Temporal change of the snubber current Is in this case can be expressed by expression (3) below:
                              I          s                =                                                            C                z                                            L                s                                              ⁢                      V            2                    ⁢          sin          ⁢                      1                                                            L                  s                                ⁢                                  C                  z                                                              ⁢          t                                    (        3        )            provided that,0≦t≦√{square root over (LsCz)}π(Effective for surge voltage absorption only during a first semi-cycle).
Here, Cz is a capacitance of the resonant capacitor, and Ls is a wiring inductance of a wiring of the snubber current path and a wiring inside the circuit elements. For example, the path of the snubber current Is in the case of FIG. 10 runs from the output point e of the rectifier diode bridge, passes though the resonant capacitor Cz, and further runs from the point g to the point f though the snubber diode Ds, a point h, and only the filter capacitor FC in the filter circuit 102, and returns to the point e through the diode bridge.
FIG. 15 illustrates temporal change of the snubber current Is flowing in the snubber circuit due to change of the wiring inductance Ls when the capacitance of the resonant capacitor Cz and the surge voltage generated by recovery in the circuit in FIG. 14 are constant. Here, where the wiring inductance Ls is L1, L2 or L3, there is a relationship of L1<L2<L3 as illustrated in FIG. 15.
In order to absorb a surge voltage generated by recovery of the rectifier diodes, it is necessary to make a large snubber current Is flow in the snubber circuit 105 immediately after generation of the voltage. It can be seen from FIG. 15 that as the wiring inductance Ls is larger, the current at t=0 when a surge voltage is generated has slower rising and thus is slower to increase. This measure can be understood from the inclination of the current waveform. Expression (4) below indicates temporal differentiation of expression (3).
                                          ⅆ                          I              s                                            ⅆ            t                          =                                            V              2                                      L              s                                ⁢          cos          ⁢                      1                                                            L                  s                                ⁢                                  C                  z                                                              ⁢          t                                    (        4        )            
It can be understood from expression (4) that as the wiring inductance Ls is larger, the inclination of the current at time t=0 when a surge voltage is generated is smaller. Also, the inclination does not depend on the capacitor capacitance Cz. In other words, as the wiring inductance Ls is larger, the surge voltage absorption deteriorates, resulting in deterioration of the function as a snubber circuit. This indicates that an overvoltage generated by recovery of the rectifier diodes and ringing of such voltage cannot be prevented and the circuit including the snubber diode Ds has no meaning. Even though the wiring inductance Ls is large and the function of the snubber circuit deteriorates, V2 does not become zero, and thus, the snubber current Is continues flowing as illustrated in FIG. 14 according to expression (3). Accordingly, an effective value of the snubber current Is is somewhat smaller in inverse proportion to a square root of the wiring inductance Ls, but the resonant capacitor Cz is charged and thus, there is no change in terms of the first problem.
A principal cause of the above problem lies in that one capacitor Cz has two functions as a resonant capacitor for making a resonant current flow in order to reduce turnoff loss of the semiconductor switches on the primary side of the transformer T, and a capacitor for a snubber circuit for absorbing a surge voltage generated when the rectifier diode bridge on the secondary side of the transformer T recovers.
In other words, the electrostatic capacitance of the capacitor Cz is determined by an amplitude of the resonant current Iz made to flow to reduce the current Iab, which is a main purpose, and a period of resonance with the resonance reactor Lz that is dominated by the leakage inductance of the transformer T. In the conventional example circuit in FIG. 10, as is clear from expression (3), the snubber current Is in the snubber path has a larger effective value as the capacitor Cz is larger. It is undesirable to make an excessive snubber current Is flow in the capacitor Cz to absorb a surge voltage. In other words, it is important that only a current for the surge absorption function is made to flow in the snubber capacitor and no extra current is made to flow in the capacitor.